Sub-gate delay adjustment using digital locked-loop

ABSTRACT

A delay locked loop (DLL) includes a delay line that delays a clock signal to generate a delayed clock signal, a phase frequency detector (PFD) for detecting a phase and/or frequency difference between the clock signal and the delayed clock signal, and a charge pump having an adjustable bias current for converting the phase and/or frequency difference taking into account a bias current adjustment into a control voltage, in which the control voltage controls an amount of delay in the delayed clock signal.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.13/629,974 filed on Sep. 28, 2012, the entirety of which is hereinincorporated by reference.

FIELD OF THE INVENTION

The present invention is generally directed to analog-to-digitalconverters (ADCs), in particular, to methods and apparatus that maycontrol time delays in clocks supplied to a continuous time sigma-delta(CTSD) modulator.

BACKGROUND INFORMATION

With the advance of semiconductor technology, the emergence of deepsub-micron or nanometer technologies allows analog designers to designwith ever faster transistors, thereby enabling the implementation ofhigh-speed circuitries and systems. In the case of a CTSD modulator, thefiner geometry (such as 65 nm) transistors allow for multi-GHz samplingclock frequency. Because of this, the input signal bandwidth that theADC can handle is also boosted dramatically, which may introduceadditional sensitivity to clock skews.

FIG. 1 is a CTSD modulator 10 which includes an input terminal 12, asummation block 14, a loop filter 16, a flash analog-to-digitalconverter (flash) 18, an optional delay locked-loop (DLL) 20, adigital-to-analog converter (DAC) 22, and an output terminal 24, wherethe summation block 14 may be part of the loop filter 16. The inputterminal 12 receives an analog signal that the CTSD modulator 10converts into a digital signal at the output terminal 24. Referring toFIG. 1, the modulator ADC 10 includes a forward signal path from theinput terminal 12 to output terminal 24, including serially connectedthe summation block 14, loop filter 16, and ADC 18, and a feedbacksignal path from the output terminal 24 to the input terminal 12,including the DAC 22. DLL 20 receives a clock signal and outputs alignedclocks to the ADC 18 and DAC 22 to drive both blocks. The summationblock 14 is configured as a subtractor that subtracts the output of DAC22 from the input signal to generate a residual signal that is fed to aninput of loop filter 16. The loop filter 16 may be a low-pass filter orbandpass filter for smoothing the residual signal which ADC 18 mayconvert into a digital output at output terminal 24. The digital outputis fed into an input of DAC 22 which converts the digital output intoanalog form to be compared with the input signal.

The two major circuit blocks, ADC 18 and DAC 22, are driven by clocksgenerated from DLL 20. During operation, the clocks fed into ADC 18 andDAC 22 may include timing differences. For example, referring to FIG. 1,ADC 18 may perform signal sampling at time instant 26, or the risingedge of a first clock cycle, while the DAC 22 may performdigital-to-analog conversion over time period 28 which starts at therising edge of a second clock cycle that follows the first clock cycle.Ideally, the time instant 26 and the outset of the time period 28 shouldhappen at the same time. However, in practice, a time difference betweenthese two may exist, which may introduce phase shifts in the frequencydomain. The phase shift may be detrimental to the stability ofhigh-order loop filters 16 contained in the feedback path.

Therefore, before the CTSD modulator is shipped to a customer, the clockskew between these two major blocks may be tuned to correct orcompensate for the high order effects. The clock to the ADC 18 may beeither delayed or advanced to match the clock to DAC 22. Delaying theclock to ADC 18 flattens the noise transfer function (NTF) of the CTSDmodulator 10, while advancing the clock would reduce the meta-stabilityof ADC 18. Therefore, it is desirable that the timing difference betweenACD 18 and DAC 22 can be fine-tuned to the degree of a few percentage ofthe clock period. For example, for certain applications, the delay needsto be below 5 ps which is finer or shorter than a simple inverter canachieve. For faster CTSD converters, the capability of sub-gate delayadjustment is very important.

Current art uses phase interpolation to generate sub-gate delays. FIG. 2illustrates a DLL that uses phase interpolation to generate sub-gatedelay. The DLL 30 includes a delay line 32 which includes a phaseinterpolator 34, a phase lock element 42 that includes a phase frequencydetector 36, a charge pump 38, and a loop filter 40, a DAC pulse driver44, and a flash clock driver 46. Additionally, DLL 30 includes aplurality of multiplexers 48, 50. To overcome sub-gate delay variationscaused by manufacturing, supplying voltage, and operating temperaturevariation (PVT), closed loop (DLL) is commonly used. Referring to FIG.2, a source clock is supplied to the delay line 32 in which the phaseinterpolation circuit 34 may generate delays between two clock outputs.The closed loop locks the input clock and the output of the dummymultiplexer 50. In practice, since DAC 22 requires a clean clock thathas little jitters, the first clock output passes through a dummymultiplexer 50 and is supplied to the phase lock element 42 at which thefirst clock output is phase locked with the source clock before suppliedto the DAC pulse driver 44. In the phase lock element 42, the phasefrequency detector 36 detects phase and frequency difference between thesource clock and the output of the dummy multiplexer 50. The charge pump38 converts the phase difference into a voltage signal that is low-passfiltered by the loop filter 40. The output from the loop filter 40 isthe control voltage that is fed back into the delay line 32 includingdelay elements. DAC driver 44 supplies a clock signal to the DAC 22. Thesecond clock output from the delay line 32 include multiple delay linesgenerated by phase interpolator 34. These delay lines are supplied tomultiplexer 48 that is controlled by a digital input. By adjusting thedigital input, the second clock output with a different amount of phasedelay is supplied to flash ADC clock driver 46 which drives the ADC 18.

In order to tap out the different phase delays generated by the phaseinterpolator 34, many levels of multiplexers are needed. The wider theadjustable range and the finer the timing resolution are, the moremultiplexers are needed. However, multiplexers need to be matched toeach other. Therefore, multiple multiplexers in CTSD modulators increasethe difficulties of circuit design.

Another challenge for nanometer circuitry is low voltage supply, whichmay be as low as 1V for certain designs.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 illustrates a continuous time sigma-delta (CTSD) modulator andits driving clocks.

FIG. 2 illustrates a delay locked-loop circuit (DLL) that usesmultiplexers to adjust clock delays.

FIG. 3 illustrates a DLL according to an embodiment of the presentinvention.

FIGS. 4A-4E illustrate charge pumps according to embodiments of thepresent invention.

FIG. 5 illustrates a detailed schematic of a charge pump according to anembodiment of the present invention.

FIGS. 6A-6B illustrate generated clock skews according to embodiments ofthe present invention.

FIG. 7 illustrates a delay element that is biased by dedicated activefilter loops according to an embodiment of the present invention.

DETAILED DESCRIPTION OF EXAMPLE EMBODIMENTS

There are needs for a CTSD modulator that includes a DLL that uses fewor no multiplexers for adjusting clock delays. Further, the DLL needs totake into consideration ease to adjust the phase in fine delayresolution (few percent of clock period), low headroom circuitry, andlow sensitivity to PVT variations.

Embodiments of the present invention may include a delay locked-loop(DLL) that includes a delay line that delays a clock signal to generatea delayed clock signal, a phase frequency detector (PFD) for detecting aphase and/or frequency difference between the clock signal and thedelayed clock signal, and a charge pump having an adjustable biascurrent for converting the phase and/or frequency difference taking intoaccount a bias current adjustment into a control voltage, in which thecontrol voltage controls an amount of delay in the delayed clock signal.

Embodiments of the present invention may include a continuous timesigma-delta (CTSD) that may include an ADC, a DAC and a DLL forgenerating clocks to drive the ADC and DAC, in which the DLL may includea delay line that delays a clock signal to generate a delayed clocksignal, a phase frequency detector (PFD) for detecting a phase and/orfrequency difference between the clock signal and the delayed clocksignal, and a charge pump having an adjustable bias current forconverting the phase and/or frequency difference taking into account abias current adjustment into a control voltage, in which the controlvoltage controls an amount of delay in the delayed clock signal.

FIG. 3 is a DLL that outputs two clock outputs whose phase differencemay be adjusted through a current input according to an embodiment ofthe present invention. The DLL 50 as shown in FIG. 3 may supply a firstclock output to a DAC in a CTSD modulator and a second clock output to aflash ADC in the CTSD, in which the phase difference between the firstand second clock outputs is conveniently adjustable through a tunablecurrent injection. Referring to FIG. 3, DLL 50 may include a delay line52, a DAC pulse driver 54, a flash ADC clock driver 56, and a phase lockelement 58 which may further include a phase frequency detector 60, acharge pump 62, and a loop filter 64. The charge pump 62 may have aninput for receiving the tunable current injection 66.

In one embodiment of the DLL 50, delay line 52 may include an inputterminal for receiving a source clock, an output terminal for outputtinga clock output that is a delayed version of the source clock, andcontrol terminals for receiving an adjustment voltage which may beadjusted to control the amount of delay in the clock output. The DACpulse driver 54 may include an input terminal for receiving the sourceclock and an output terminal for supplying the source clock to the DACand to a first input terminal of the phase frequency detector 60. Theflash ADC clock driver 56 may include an input terminal coupled to theoutput terminal of the delay line 52 for receiving the delayed sourceclock and an output terminal for supplying the delayed source clock tothe flash ADC and to a second input terminal of the phase frequencydetector 60. The phase frequency detector 60 may detect a phase and/orfrequency difference between the signals at the first and secondterminals and output the phase and/or frequency difference to the chargepump 62. The charge pump 62 may also include an additional control inputterminal for receiving the tunable current injection 66 to insert anoffset current intended by a user. The charge pump 62 may convert thephase and/or frequency difference into a voltage signal indicating thedifference taking into consideration of the tunable current injection66. By intentionally injecting current into the charge pump 62, a phaseskew may be inserted into the clock to the flash ADC. The loop filter 64may smooth the voltage output from the charge pump 62 and supply thesmoothed output to the control terminal of the delay line 52.

In operation, if there is no current injection to the charge pump, theclock to the DAC and the clock to the flash ADC are phase-locked throughthe feedback loop to the control terminal of the delay line 52. However,if there is current injection to the charge pump, a phase skew betweenthe clock to the DAC and the clock to the flash ADC may be intentionallygenerated. Thus, by controlling the amount of the tunable currentinjection 66 at the charge pump 62, the phase skew between the clock tothe DAC and the clock to the flash ADC may be adjusted.

Embodiments of the present invention as shown in FIG. 3 may control thephase skew through current injection without the need for a phaseinterpolator and multiplexers. Further, embodiments of the presentinvention directly compare the timing difference between the two clocksto the DAC and the flash ADC, thereby reducing the DLL's sensitivity toPVT.

Embodiments of the present invention may include a PFD 60 that mayinclude two dynamic latches and a feedback AND gate that may reset thetwo latches. When the two latches are locked in a steady state, the PFDmay output an up (UP) pulse at an output of the first latch and a down(DN) pulse at an output of the second latch. If there is no currentmismatch between the bias currents of the UP and DN branches, the UP andDN pulses match, or rise at the same time and have the same pulse width.However, if there is current mismatch, the UP and DN pulses may beskewed and may have different pulse widths. The charge pump 68 maytranslate the problem of matching the timing of UP and DN pulses into acurrent matching problem.

Embodiments of the present invention may include different forms ofcharge pumps that may convert the timing mismatches into a currentmatching problem. FIG. 4A is a charge pump 68 according to an embodimentof the present invention. As shown in FIG. 4A, the charge pump 68 mayinclude a pair of MOS transistors 70.1, 70.2, a pair of resistors 72.1,72.2, a first pair of switches 74.1, 74.2, a second pair of switches76.1, 76.2, a pair of current sources 78.1, 78.2, and output nodes 80.1,80.2. In one embodiment, the pair of MOS transistors may be PMOStransistors whose sources may be coupled to a voltage supply 81, whosegates may be coupled together, and whose drains may be coupled to theoutput nodes 80.1, 80.2. Resistor 72.1 may be coupled between the gateof PMOS 70.1 and the output node 80.1, and resistor 72.2 may be coupledbetween the gate of PMOS 70.2 and the output node 80.2. The firstswitches 74.1, 74.2 respectively may be coupled between current source78.1 and output nodes 80.1, 80.2, respectively. The first pair ofswitches may be controlled by the UP pulse from PFD 60. The UP pulsetriggers switch 74.1 and the inverted UP pulse triggers switch 74.2. Thesecond pair of switches may be controlled by the DN pulse from the PFD60. The DN pulse triggers switch 76.2 and the inverted DN pulse triggersswitch 76.1.

According to the embodiment as shown in FIG. 4A, the charge pump 68 mayaccommodate low voltage supply 81. One known approach to allowing fastswitching frequency is H-bridge technology. Unfortunately, H-bridgetechnology suffers from the problem of insufficient headroom. In theembodiment as shown in FIG. 4A, the charge pump 68 uses resistors 72.1,72.2 across the output nodes 80.1, 80.2 to provide DC bias for PMOStransistors 70.1, 70.2. This self-biased scheme removes the concern forcurrent matching between NMOS and PMOS current sources as used in theH-bridge technology.

FIGS. 4D and 4E show further embodiments of a charge pump according tothe present invention. The charge pump 440 shown in FIG. 4D may be ageneralized version of the charge pump 430 shown in FIG. 4C. The chargepump 450 shown in FIG. 4E may be a generalized version of the chargepump 410, 420 shown in FIGS. 4A and 4B. The components of each of chargepumps 440 and 450 are the same and have the same functions ascorresponding components described with respect to FIG. 4A, unlessotherwise described herein. Each of the charge pumps 440 and 450 mayinclude a first portion, which may include one or more fixed currentcomponents 70.1, 70.2 with common mode bias circuits. Each of chargepumps 440 and 450 may further include a second portion, which mayinclude switching structures 74 and 76 and tunable offset currents I±The first portion and the second portion may be joined by and coupledto output nodes 80.1, 80.2. The second portion may be directly orswitchably connected to the first portion. For example, all or some ofthe dashed switching paths may be replaced by direct connections withoutswitches such that the tunable offset currents are directly connected tothe output nodes, e.g., without switches or with switches always turnedon.

Since the supply is low, it is preferable that the output voltages donot swing much. Also, the output nodes may need to have equal potentialto minimize unwanted offsets caused by current mismatches in charge pump68. In one embodiment, outputs of the charge pump 68 may be coupled toan active loop filter (not shown), which may clamp voltages at theoutput nodes 80.1, 80.2 and keep them at the same voltage level (if thegain of the active filter is large enough). In one embodiment as shownin FIG. 4A, the self-biased PMOS transistors are HVT (high-thresholdvoltage) devices which may provide good DC common mode input voltage forthe active loop filter, whose input pair may be designed to be LVT(low-threshold voltage) PMOS. In this way, enough headroom may becreated.

In one embodiment, current sources 78.1, 78.2 may be constructed usingNMOS current mirrors. When there are mismatches between current mirrors,the UP and DN pulses may be skewed because the active loop filter is anintegrator and in steady state, has no net current flowing out of theoutput nodes during one clock cycle for the charge pump. This mechanismmay be utilized here to purposely generate an intended skew between theUP and DN pulses. The intended time skew may be reflected as a currentadjustment ΔI. Here, ΔI may be the tunable current injection 66 that maybe adjusted either automatically or manually by a user.

Embodiments of the present invention may include various charge pumpsthat include adjustable bias currents. For example, there may bedifferent circuitries to connect the voltage supply 81 to the outputnodes 80.1, 80.2. FIG. 4B is a charge pump according to an alternativeembodiment of the present invention that includes current sources 75.1,75.2 that connects voltage supply 81 to output nodes 80.1, 80.2. Asshown in FIG. 4B, MOS transistors 70.1, 70.2 of FIG. 4A may be replacedby current sources 75.1, 75.2 that provide fixed current to the outputnodes 80.1, 80.2 and resistors 72.1, 72.2 of FIG. 4A may be replaced bya common mode feedback circuit (CMFB) 73 that may have a first inputcoupled to the first output node 80.1 and a second input coupled to thesecond output node 80.2. In one embodiment, CMFB 73 may be the resistorpair 72.1, 72.2, or alternatively, CMFB may be an amplifier that outputsthe difference signal between its two input terminals. The output ofCMFB 73 may be coupled to the control terminals of the current sources75.1, 75.2 of FIG. 4B. The current sources 75.1, 75.2 may be the PMOSdevices as shown in FIG. 4A, or alternatively, NMOS devices. FIG. 4C isa charge pump according to another alternative embodiment of the presentinvention to construct the bottom half of the circuit. As shown in FIG.4C, the current sources 78.1, 78.2 may be split into two parts in whichthe first part may include the current source I and a second part ΔI.The first part (I) of the current sources 78.1, 78.2 may be switchablyconnected to the first and second output nodes via switches 74.1, 74.2,76.1, 76.2. Additionally, the second part (Δ) of the current source maybe directly coupled to the first and second output nodes, respectively.

The current from the NMOS current mirrors may be divided into multiplebranches, in which the current direction of each branch may be steeredby a cascode switching pair. As shown in FIG. 5, the charge pump 82 mayinclude NMOS switches 84.1, 84.2 controlled by UP pulses and NMOSswitches 86.1, 86.2 controlled by DN pulses. FIG. 5 is a charge pump 82that includes multiple branches of current mirrors according to anembodiment of the present invention. The current sources of the chargepump may be a plurality of current mirror branches that are connected inparallel. Each branch may include a large NMOS bias device and a pair ofNMOS cascode switching devices, as in this embodiment, NMOS transistors88.1, 90.1, 92.1 that may form a first current branch that includes afirst leg (NMOS 88.1) that steers a current to the switches 84.1, 84.2and a second leg (NMOS 90.1) that steers a current to the switches 86.1,86.2. The sources of NMOS 88.1, 90.1 may be coupled to NMOS 92.1. Thecascode switching pair in each branch may be independently digitallycontrolled. In this embodiment, the gates of NMOS 88.1, 90.1 may bedigitally controlled by a digital code B, to be connected to either acascode bias voltage or a turn-off voltage. Based on the digital code B,the direction of this single current branch may be selected so that thecurrent may be drained from either UP or DN branches, thus creating atunable current injection to the DLL. In this way, a DAC is in essenceincorporated into the charge pump.

The purpose of having the DAC is to translate the timing problem into acurrent matching problem. The minimum pulse widths resulting from thepropagation delays in the PFD 60 are dependent on process voltagetemperature (PVT). If the offset current is kept constant, theintentionally generated skews depend on PVT, which is undesirable. FIG.6A illustrates such a PVT dependency for the skew. The reason is thatthe switching of the charge pump is coupled with the offset current.

Measures may be taken to decouple the switching of the charge pump andthe offset current. In one embodiment, a signal generator (not shown)may generate a clock signal to activate or deactivate the charge pumpoffset current. FIG. 6B shows that the offset current (the tunablecurrent injection ΔI) is turned off during UP and DN pulses. As shown inFIG. 6B, the UP and DN pulses may occur during the time when the chargepump DAC is deactivated, which may greatly mitigate the PVT dependency.

The delay line 52 may include both PMOS and NMOS devices that arecurrent starved to maximize the tuning range for the offset current.Current art uses one simple bias scheme to provide biases to both PMOSand NMOS devices. In one embodiment, the PMOS and NMOS devices of thedelay line 52 may respectively have dedicated bias. In this way, thePMOS devices and the NMOS devices may have substantially equal amount ofwork load. FIG. 7 is a delay line that has respective dedicated biasesfor PMOS devices and NMOS devices according to an embodiment of thepresent invention. As shown in FIG. 7, a primary loop provides a firstdedicate bias to all of the PMOS devices in the delay line 52, and asecondary loop provides a second dedicated bias to all of the NMOSdevices in the delay line 52. In this way, the PMOS and NMOS devices mayhave separate biases.

Those skilled in the art may appreciate from the foregoing descriptionthat the present invention may be implemented in a variety of forms, andthat the various embodiments may be implemented alone or in combination.Therefore, while the embodiments of the present invention have beendescribed in connection with particular examples thereof, the true scopeof the embodiments and/or methods of the present invention should not beso limited since other modifications will become apparent to the skilledpractitioner upon a study of the drawings, specification, and followingclaims.

What is claimed is:
 1. A charge pump for converting at least one of: aphase and a frequency difference into a control voltage, the charge pumpcomprising: a first current component coupled between a voltage supplyand a first output node; a second current component coupled between thevoltage supply and a second output node, a control node of the firstcurrent component being coupled to a control node of the second currentcomponent; a common mode feedback circuit including a first inputcoupled to the first output node, a second input coupled to the secondoutput node, and an output coupled to the control nodes of the first andsecond current components; wherein a first bias current source suppliesa first tunable bias current directly to the first output node andswitchably supplies a first constant bias current to the first andsecond output nodes, and a second bias current source supplies a secondtunable bias current directly to the first output node and switchablysupplies a second constant bias current to the first and second outputnodes.
 2. The charge pump of claim 1, wherein the first and secondtunable bias currents are of a substantially same magnitude butsubstantially opposite directions.
 3. The charge pump of claim 1,wherein the first and second current components include one of a p-typeMOSFET (PMOS) and an n-type MOSFET (NMOS).
 4. The charge pump of claim1, wherein the common mode feedback circuit includes: a first resistorcoupled between the first output node and a gate of the first currentcomponent; and a second resistor coupled between the second output nodeand a gate of the second current component.
 5. The charge pump of claim1, wherein the common mode feedback circuit includes an amplifier. 6.The charge pump of claim 1, wherein the first and second bias currentsources are connected directly to the output nodes.
 7. A charge pump forconverting a phase and/or frequency difference into a control voltage,the charge pump comprising: a first current component coupled between avoltage supply and a first output node; a second current componentcoupled between the voltage supply and a second output node, a controlnode of the first current source component being coupled to a controlnode of the second current source component; and a common mode feedbackcircuit including a first input coupled to the first output node, asecond input coupled to the second output node, and an output coupled tothe control nodes of the first and second current source components.wherein a first bias current source supplies a first adjustable biascurrent to the first and second output nodes, and a second bias currentsource supplies a second adjustable bias current to the first and secondoutput nodes.
 8. The charge pump of claim 7, further comprising: a firstswitch coupled between the first bias current source and the firstoutput node; a second switch coupled between the first bias currentsource and the second output node; a third switch coupled between thesecond bias current source and the first output node; and a fourthswitch coupled between the second bias current source and the secondoutput node.
 9. The charge pump of claim 7, wherein the first offsetcurrent is turned off before the first pulse occurs, the second offsetcurrent is turned off before the second pulse occurs, and the first andsecond offset currents are tunable only when both the first and secondpulses are off.
 10. The charge pump of claim 7, wherein the first andsecond current source components include one of a p-type MOSFET (PMOS)and an n-type MOSFET (NMOS).
 11. The charge pump of claim 7, wherein thecommon mode feedback circuit includes an amplifier.
 12. The charge pumpof claim 7, wherein the common mode feedback circuit includes: a firstresistor coupled between the first output node and a gate of the firstcurrent component; and a second resistor coupled between the secondoutput node and a gate of the second current component.
 13. A chargepump comprising: a fixed current component coupled between a voltagesupply and a pair of output nodes; a common mode bias circuit coupled tothe fixed current component and the pair of output nodes; a switchingstructure coupled to the output nodes; and a pair of tunable offsetcurrent sources coupled to the switching structure.
 14. The charge pumpof claim 13, wherein the pair of tunable offset current sourcesswitchably supplies a bias current to the pair of output nodes.
 15. Thecharge pump of claim 13, wherein the pair of tunable offset currentsources is directly coupled to the output nodes.